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Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
DESCRIPTION
The NE5210 is a 7k transimpedance wide band, low noise amplifier with differential outputs, particularly suitable for signal recovery in fiber-optic receivers. The part is ideally suited for many other RF applications as a general purpose gain block.
PIN CONFIGURATION
D Package
GND2 GND2 1 2 3 4 5 6 7 14 13 12 11 10 9 8 OUT (-) GND2 OUT (+) GND1 GND1 GND1 GND1
FEATURES
NC IIN NC VCC1 VCC2
* Low noise: 3.5pA/Hz * Single 5V supply * Large bandwidth: 280MHz * Differential outputs * Low input/output impedances * High power supply rejection ratio * High overload threshold current * Wide dynamic range * 7k differential transresistance
APPLICATIONS
TOP VIEW
SD00318
* Fiber-optic receivers, analog and digital * Current-to-voltage converters
ORDERING INFORMATION
DESCRIPTION 14-Pin Plastic Small Outline (SO) Package
* Wideband gain block * Medical and scientific instrumentation * Sensor preamplifiers * Single-ended to differential conversion * Low noise RF amplifiers * RF signal processing
TEMPERATURE RANGE 0 to +70C
ORDER CODE NE5210D
DWG # SOT108-1
ABSOLUTE MAXIMUM RATINGS
SYMBOL VCC TA TJ TSTG PDMAX IINMAX Power supply Operating ambient temperature range Operating junction temperature range Storage temperature range Power dissipation, TA=25C (still air)1 Maximum input current2 PARAMETER RATING 6 0 to +70 -55 to +150 -65 to +150 1.0 5 UNIT V C C C W mA
NOTES: 1. Maximum dissipation is determined by the operating ambient temperature and the thermal resistance: JA=125C/W. 2. The use of a pull-up resistor to VCC for the PIN diode, is recommended.
RECOMMENDED OPERATING CONDITIONS
SYMBOL VCC TA TJ Supply voltage Ambient temperature range Junction temperature range PARAMETER RATING 4.5 to 5.5 0 to +70 0 to +90 UNIT V C C
1995 Apr 26
1
853-1654 15170
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
DC ELECTRICAL CHARACTERISTICS
Min and Max limits apply over operating temperature range at VCC=5V, unless otherwise specified. Typical data applies at VCC=5V and TA=25C. SYMBOL VIN VO VOS ICC IOMAX IIN IINMAX PARAMETER Input bias voltage Output bias voltage Output offset voltage Supply current Output sink/source current1 Input current (2% linearity) Maximum input current overload threshold Test Circuit 8, Procedure 2 Test Circuit 8, Procedure 4 21 3 120 160 TEST CONDITIONS LIMITS Min 0.6 2.8 Typ 0.8 3.3 0 26 4 160 240 Max 0.95 3.7 80 32 UNIT V V mV mA mA A A
NOTES: 1. Test condition: output quiescent voltage variation is less than 100mV for 3mA load current.
AC ELECTRICAL CHARACTERISTICS
Typical data and Min/Max limits apply at VCC=5V and TA=25C. SYMBOL RT RO RT RO f3dB RIN CIN R/V R/T IN PARAMETER Transresistance (differential output) Output resistance (differential output) Transresistance (single-ended output) Output resistance (single-ended output) Bandwidth (-3dB) Input resistance Input capacitance Transresistance power supply sensitivity Transresistance ambient temperature sensitivity RMS noise current spectral density (referred to input) Integrated RMS noise current over the bandwidth (referred to input) CS=01 VCC=50.5V TA=TA MAX-TA MIN f=10MHz, TA=25C Test Circuit 2 TA=25C Test Circuit 2 f=100MHz f=200MHz f=300MHz f=100MHz CS=1pF Power supply rejection ratio2 (VCC1=VCC2) Power supply rejection ratio2 (VCC1) Power supply rejection ratio2 (VCC2) Power supply rejection configuration) ratio2 (ECL f=200MHz f=300MHz PSRR PSRR PSRR PSRR DC tested, VCC=0.1V Equivalent AC test circuit 3 DC tested, VCC=0.1V Equivalent AC test circuit 4 DC tested, VCC=0.1V Equivalent AC test circuit 5 f=0.1MHz, Test Circuit 6 20 20 37 56 71 40 66 89 36 36 65 23 dB dB dB dB nA TEST CONDITIONS DC tested, RL= Test Circuit 8, Procedure 1 DC tested DC tested, RL= DC tested Test Circuit 1, TA=25C LIMITS Min 4.9 16 2.45 8 200 Typ 7 30 3.5 15 280 60 7.5 9.6 0.05 3.5 20 0.1 6 Max 10 42 5 21 UNIT k k MHz pF %/V %/C pA/Hz
IT
1995 Apr 26
2
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
AC ELECTRICAL CHARACTERISTICS (Continued)
SYMBOL VOMAX VINMAX tR PARAMETER Maximum output voltage swing differential Maximum input amplitude for output duty cycle of 505%3 Rise time for 50 mVP-P output signal4 TEST CONDITIONS RL= Test Circuit 8, Procedure 3 Test Circuit 7 Test Circuit 7 LIMITS Min 2.4 650 0.8 1.2 Typ 3.2 Max UNIT VP-P mVP-P ns
NOTES: 1. Package parasitic capacitance amounts to about 0.2pF 2. PSRR is output referenced and is circuit board layout dependent at higher frequencies. For best performance use RF filter in VCC line. 3. Guaranteed by linearity and overload tests. 4. tR defined as 20-80% rise time. It is guaranteed by a -3dB bandwidth test.
TEST CIRCUITS
SINGLE-ENDED NETWORK ANALYZER RT [ V OUT V IN R + 2 @ S21 @ R RT + V OUT V IN R + 4 @ S21 @ R DIFFERENTIAL
S-PARAMETER TEST SET PORT 1 5V VCC1 VCC2 33 0.1F ZO = 50 PORT 2
RO [ ZO
1 ) S22 * 33 1 * S22
R O + 2Z O
1 ) S22 * 66 1 * S22
0.1F ZO = 50
OUT R = 1k IN DUT
33 OUT 50 GND1 GND2
0.1F
RL = 50
Test Circuit 1
SPECTRUM ANALYZER
5V VCC1 VCC2 33 AV = 60DB 0.1F ZO = 50
OUT NC IN DUT
33 OUT
0.1F
RL = 50 GND1 GND2
Test Circuit 2
SD00319
1995 Apr 26
3
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
NETWORK ANALYZER
5V 10F 0.1F PORT 1
S-PARAMETER TEST SET PORT 2
10F 0.1F 16
CURRENT PROBE 1mV/mA
CAL
VCC1
VCC2 OUT
33
0.1F 50 TEST 100 BAL. 0.1F TRANSFORMER NH0300HB
IN 33 OUT
UNBAL.
GND1
GND2
Test Circuit 3
NETWORK ANALYZER
5V 10F 0.1F PORT 1
S-PARAMETER TEST SET PORT 2
10F 0.1F 16
CURRENT PROBE 1mV/mA
CAL
5V
10F 0.1F IN
VCC2
VCC1 OUT
33
0.1F 50 TEST 100 BAL. 0.1F TRANSFORMER NH0300HB
UNBAL.
33 OUT GND1 GND2
Test Circuit 4
SD00320
1995 Apr 26
4
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
NETWORK ANALYZER
5V 10F 0.1F PORT 1
S-PARAMETER TEST SET PORT 2
10F 0.1F 16
CURRENT PROBE 1mV/mA
CAL
5V
10F 0.1F IN
VCC1
VCC2 OUT
33
0.1F 50 TEST 100 BAL. 0.1F TRANSFORMER NH0300HB
UNBAL.
33 OUT GND1 GND2
Test Circuit 5
NETWORK ANALYZER
S-PARAMETER TEST SET GND PORT 1 PORT 2
10F 0.1F 16
CURRENT PROBE 1mV/mA
CAL
GND1
GND2 OUT
33
0.1F 50 TEST 100 BAL. 0.1F TRANSFORMER NH0300HB
IN 33 OUT 5.2V 10F 0.1F VCC1 VCC2
UNBAL.
Test Circuit 6
SD00321
1995 Apr 26
5
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
PULSE GEN.
VCC1
VCC2 33 0.1F A 33 OUT 0.1F ZO = 50 OSCILLOSCOPE B ZO = 50
0.1F 1k IN DUT
OUT
50 GND1 GND2 Measurement done using differential wave forms
Test Circuit 7
SD00322
1995 Apr 26
6
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TEST CIRCUITS (Continued)
Typical Differential Output Voltage vs Current Input 5V
OUT + IN IIN (A) GND1 GND2 DUT OUT -
+ VOUT (V) -
2.00 1.60 DIFFERENTIAL OUTPUT VOLTAGE (V) 1.20 0.80 0.40 0.00 -0.40 -0.80 -1.20 -1.60 -2.00 -400 -320 -240 -160 -80 0 80 160 240 320 400
CURRENT INPUT (A)
NE5210 TEST CONDITIONS Procedure 1 RT measured at 60A RT = (VO1 - VO2)/(+60A - (-60A)) Where: VO1 Measured at IIN = +60A VO2 Measured at IIN = -60A Procedure 2 Linearity = 1 - ABS((VOA - VOB) / (VO3 - VO4)) Where: VO3 Measured at IIN = +120A VO4 Measured at IIN = -120A V + R T @ () 120mA) ) V OA OB V + R T @ (* 120mA) ) V OB OB
Procedure 3
VOMAX = VO7 - VO8 Where: VO7 Measured at IIN = +260A VO8 Measured at IIN = -260A
Procedure 4
IIN Test Pass Conditions: VO7 - VO5 > 20mV and V06 - VO5 > 20mV Where: VO5 Measured at IIN = +160A VO6 Measured at IIN = -160A VO7 Measured at IIN = +260A VO8 Measured at IIN = -260A
Test Circuit 8
SD00323
1995 Apr 26
7
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS
NE5210 Supply Current vs Temperature
32 TOTAL SUPPLY CURRENT (mA) (I CC1+ I CC2) OUTPUT BIAS VOLTAGE (V) 30 28 26 24 22 20 18 -10 0
NE5210 Output Bias Voltage vs Temperature
3.50 VCC = 5.0V OUTPUT VOLTAGE (V) 3.46 4.5
Output Voltage vs Input Current
+25C +85C +125C -55C 3.0
3.42
PIN 14 PIN 12
3.38
+125C
3.34
10 20 30 40 50 60 70
80
3.30 -10 0
10 20 30 40 50 60 70
80
2.5 -300.0
+85C 0 INPUT CURRENT (A) +300.0
AMBIENT TEMPERATURE (C)
AMBIENT TEMPERATURE (C)
NE5210 Input Bias Voltage vs Temperature
900 OUTPUT BIAS VOLTAGE (V) 5.5V INPUT BIAS VOLTAGE (mV) 850 5.0V 4.5V 800
NE5210 Output Bias Voltage vs Temperature
DIFFERENTIAL OUTPUT VOLTAGE (V) 4.1 PIN 14 3.9 3.7 3.5 3.3 3.1 2.9 2.7 -10 0 10 20 30 40 50 60 70 80 4.5V 5.0V 5.5V
Differential Output Voltage vs Input Current
2.0 5.0V 4.5V 5.5V
0
750
4.5V 5.0V 5.5V 0 INPUT CURRENT (A) +300.0
700 -10 0
10 20 30 40 50 60 70
80
-2.0 -300.0
AMBIENT TEMPERATURE (C)
AMBIENT TEMPERATURE (C)
NE5210 Output Offset Voltage vs Temperature
20
NE5210 Differential Output Swing vs Temperature
4.0 3.8 3.6 5.5V 3.4 3.2 5.0V 3.0 2.8 2.6 2.4 2.2 -10 0 10 20 30 40 50 60 70 80 4.5V DC TESTED RL = DIFFERENTIAL OUTPUT VOLTAGE (V)
Differential Output Voltage vs Input Current
2.0
VOS = VOUT12 - VOUT14 0 4.5V -20 5.0V -40 5.5V -60
DIFFERENTIAL OUTPUT SWING (V)
OUTPUT OFFSET VOLTAGE (mV)
0
-55C +25C +85C +125C 0 INPUT CURRENT (A) +300.0
-80 -10 0
10 20 30 40 50 60 70
80
-2.0 -300.0
AMBIENT TEMPERATURE (C)
AMBIENT TEMPERATURE (C)
SD00324
1995 Apr 26
8
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
Gain vs Frequency
8 7 6 GAIN (dB) 5 4 3 2 1 0 -1 1 10 100 FREQUENCY (MHz) 1000 4.5V 5.0V 5.5V PIN 12 VCC = 5V RL = 50 GAIN (dB) 8 7 6 5 4 3 2 1 0 -1 1 10 100 FREQUENCY (MHz) 1000 4.5V 5.0V 5.5V
Gain vs Frequency
PIN 12 VCC = 5V RL = 50
DIFFERENTIAL TRANSRESISTANCE (k )
NE5210 Differential Transresistance vs Temperature
8.6 8.4 8.2 8.0 7.8 7.6 5.5V 5.0V 4.5V RL =
7.4 -10 0
10 20 30 40 50 60 70
80
AMBIENT TEMPERATURE (C)
Gain vs Frequency
8 7 6 GAIN (dB) 5 4 3 2 1 0 -1 1 10 100 FREQUENCY (MHz) 1000 +85C +125C 25C +125C -55C PIN 12 VCC = 5V GAIN (dB) -55C 8 7 6 5 4 3 2 1 0 -1 1
Gain vs Frequency
NE5210 Bandwidth vs Temperature
450 PIN 14 VCC = 5V -55C 400 5.5V 350 5.0V 300 250 200 -10 0 4.5V PIN 12 SINGLE-ENDED RL =
85C
25C
+125C 10 100 FREQUENCY (MHz) 1000
BANDWIDTH (MHz)
10 20 30 40 50 60 70
80
AMBIENT TEMPERATURE (C)
Gain and Phase Shift vs Frequency
8 7 6 GAIN (dB) 5 4 3 2 1 0 -1 1 10 100 FREQUENCY (MHz) 1000 -180 0 PIN 12 VCC = 5V TA = 25C 180 8 7 90 PHASE ( o ) GAIN (dB) 6 5 4 3 2 1 0 -1 1
Gain and Phase Shift vs Frequency
360 PIN 14 VCC = 5V TA = 25C 50 40 30 20 10 0 10 100 FREQUENCY (MHz) 1000
NE5210 Typical Bandwidth Distribution (70 Parts from 4 Wafer Lots)
PIN 12 SINGLE-ENDED RL = 50 VCC = 5.0V TA = 25C
180
-90
90
0 223 255 287 319 351 FREQUENCY (MHz) 383
POPULATION (%)
270 PHASE ( o )
SD00325
1995 Apr 26
9
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
TYPICAL PERFORMANCE CHARACTERISTICS (Continued)
NE5210 Output Resistance vs Temperature
17 OUTPUT RESISTANCE ( ) OUTPUT RESISTANCE ( ) VCC = 5.0V DC TESTED 16 PIN 14 ROUT 5.0V
NE5210 Output Resistance vs Temperature
16 OUTPUT RESISTANCE ( ) PIN 12 OUTPUT REFERRED 15 4.5V 14 5.0V 5.5V 13 17
NE5210 Output Resistance vs Temperature
PIN 14 OUTPUT REFERRED 16 4.5V 15 5.0V 5.5V 14
15
14 5.0V 13 PIN 12 ROUT
12 -10 0 10 20 30 40 50 60 70 80 AMBIENT TEMPERATURE (C)
12 -10 0 10 20 30 40 50 60 70 80 AMBIENT TEMPERATURE (C)
13 -10 0 10 20 30 40 50 60 70 80 AMBIENT TEMPERATURE (C)
Output Resistance vs Frequency
NE5210 Power Supply Rejection Ratio vs Temperature
POWER SUPPLY REJECTION RATIO (dB) 40 39 38 37 36 35 34 33 -10 0 10 20 30 40 50 60 70 80 AMBIENT TEMPERATURE (C) 0.1 20 40 VCC1 = VCC2 = 5.0V VCC = 0.1V DC TESTED OUTPUT REFERRED 10 8 6 DELAY (ns) 4 2 0
Group Delay
OUTPUT RESISTANCE ( )
80 70 60 50 40 30 20 10 0 0.1 1 10 FREQUENCY (MHz) PIN 14 100 200 PIN 12 VCC = 5.0V TA = 25C
VCC = 5V TA = 25C
60 80 100 120 140 160 180 200 FREQUENCY (MHz)
Output Step Response
VCC = 5V TA = 25C 20mV/Div
0
2
4
6
8
10 (ns)
12
14
16
18
20
SD00326
1995 Apr 26
10
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
THEORY OF OPERATION
Transimpedance amplifiers have been widely used as the preamplifier in fiber-optic receivers. The NE5210 is a wide bandwidth (typically 280MHz) transimpedance amplifier designed primarily for input currents requiring a large dynamic range, such as those produced by a laser diode. The maximum input current before output stage clipping occurs at typically 240A. The NE5210 is a bipolar transimpedance amplifier which is current driven at the input and generates a differential voltage signal at the outputs. The forward transfer function is therefore a ratio of the differential output voltage to a given input current with the dimensions of ohms. The main feature of this amplifier is a wideband, low-noise input stage which is desensitized to photodiode capacitance variations. When connected to a photodiode of a few picoFarads, the frequency response will not be degraded significantly. Except for the input stage, the entire signal path is differential to provide improved power-supply rejection and ease of interface to ECL type circuitry. A block diagram of the circuit is shown in Figure 1. The input stage (A1) employs shunt-series feedback to stabilize the current gain of the amplifier. The transresistance of the amplifier from the current source to the emitter of Q3 is approximately the value of the feedback resistor, RF=3.6k. The gain from the second stage (A2) and emitter followers (A3 and A4) is about two. Therefore, the differential transresistance of the entire amplifier, RT is RT + V OUT(diff) + 2R F + 2(3.6K) + 7.2kW I IN
OUTPUT + A3
INPUT A1 A2
RF
A4 OUTPUT -
SD00327
Figure 1. NE5210 - Block Diagram
BANDWIDTH CALCULATIONS
The input stage, shown in Figure 3, employs shunt-series feedback to stabilize the current gain of the amplifier. A simplified analysis can determine the performance of the amplifier. The equivalent input capacitance, CIN, in parallel with the source, IS, is approximately 7.5pF, assuming that CS=0 where CS is the external source capacitance. Since the input is driven by a current source the input must have a low input resistance. The input resistance, RIN, is the ratio of the incremental input voltage, VIN, to the corresponding input current, IIN and can be calculated as: V RF + 3.6K + 51W R IN + IN + 71 I IN 1 ) A VOL More exact calculations would yield a higher value of 60. Thus CIN and RIN will form the dominant pole of the entire amplifier; f *3dB + 2p 1 R IN C IN
The single-ended transresistance of the amplifier is typically 3.6k. The simplified schematic in Figure 2 shows how an input current is converted to a differential output voltage. The amplifier has a single input for current which is referenced to Ground 1. An input current from a laser diode, for example, will be converted into a voltage by the feedback resistor RF. The transistor Q1 provides most of the open loop gain of the circuit, AVOL70. The emitter follower Q2 minimizes loading on Q1. The transistor Q4, resistor R7, and VB1 provide level shifting and interface with the Q15 - Q16 differential pair of the second stage which is biased with an internal reference, VB2. The differential outputs are derived from emitter followers Q11 - Q12 which are biased by constant current sources. The collectors of Q11 - Q12 are bonded to an external pin, VCC2, in order to reduce the feedback to the input stage. The output impedance is about 17 single-ended. For ease of performance evaluation, a 33 resistor is used in series with each output to match to a 50 test system.
Assuming typical values for RF = 3.6k, RIN = 60, CIN = 7.5pF f *3dB + 2p 1 + 354MHz 7.5pF 60
VCC1 VCC2 R1 Q2 Q1 R2 GND1 PHOTODIODE R5 R4 GND2 R7 VB2 Q3 R3 Q4 + Q15 R14 Q16 R15 + OUT+ R12 R13 Q11 Q12 OUT-
INPUT
SD00328
Figure 2. Transimpedance Amplifier
1995 Apr 26
11
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
VCC R1 INPUT IIN IB Q1 R2 VIN IF VEQ3 IC1 Q2 Q3 R3
For a given wavelength ; (meters) Energy of one Photon = hc watt sec (Joule) l Where h=Planck's Constant = 6.6 x 10-34 Joule sec. c = speed of light = 3 x 108 m/sec c / = optical frequency (Hz) No. of incident photons/sec= where P=optical incident power P No. of incident photons/sec = hs l where P = optical incident power
RF R4
P No. of generated electrons/sec = h @ hs l
SD00329
where = quantum efficiency + no. of generated electron hole paris no. of incident photons
Figure 3. Shunt-Series Input Stage The operating point of Q1, Figure 2, has been optimized for the lowest current noise without introducing a second dominant pole in the pass-band. All poles associated with subsequent stages have been kept at sufficiently high enough frequencies to yield an overall single pole response. Although wider bandwidths have been achieved by using a cascode input stage configuration, the present solution has the advantage of a very uniform, highly desensitized frequency response because the Miller effect dominates over the external photodiode and stray capacitances. For example, assuming a source capacitance of 1pF, input stage voltage gain of 70, RIN = 60 then the total input capacitance, CIN = (1+7.5) pF which will lead to only a 12% bandwidth reduction.
P hs @ e Amps (Coulombs sec.) N I+h @ l where e = electron charge = 1.6 x 10-19 Coulombs h @e Responsivity R = hs Amp/watt l I + P@R Assuming a data rate of 400 Mbaud (Bandwidth, B=200MHz), the noise parameter Z may be calculated as:1 Z+ I EQ 66 @ 10 *9 + + 2063 qB (1.6 @ 10 *19)(200 @ 10 6)
NOISE
Most of the currently installed fiber-optic systems use non-coherent transmission and detect incident optical power. Therefore, receiver noise performance becomes very important. The input stage achieves a low input referred noise current (spectral density) of 3.5pA/Hz. The transresistance configuration assures that the external high value bias resistors often required for photodiode biasing will not contribute to the total noise system noise. The equivalent input RMS noise current is strongly determined by the quiescent current of Q1, the feedback resistor RF, and the bandwidth; however, it is not dependent upon the internal Miller-capacitance. The measured wideband noise was 66nARMS in a 200MHz bandwidth.
where Z is the ratio of RMS noise output to the peak response to a single hole-electron pair. Assuming 100% photodetector quantum efficiency, half mark/half space digital transmission, 850nm lightwave and using Gaussian approximation, the minimum required optical power to achieve 10-9 BER is: P avMIN + 12 hc B Z + 12 2.3 @ 10 *19 l 200 @ 10 6 2063 + 1139nW + * 29.4dBm where h is Planck's Constant, c is the speed of light, is the wavelength. The minimum input current to the NE5210, at this input power is: I avMIN + qP avMIN l hc
*9 *19 + 1139 @ 10 @ 1.6 @ 10 2.3 @ 10 *19 = 792nA
DYNAMIC RANGE CALCULATIONS
The electrical dynamic range can be defined as the ratio of maximum input current to the peak noise current: Electrical dynamic range, DE, in a 200MHz bandwidth assuming IINMAX = 240A and a wideband noise of IEQ=66nARMS for an external source capacitance of CS = 1pF. D E + 20log (Max. input current) (PK) (Peak noise current) (RMS) @ 2 (240 @ 10 *6) ( 2 66 10 *9) + 68dB
Choosing the maximum peak overload current of IavMAX=240A, the maximum mean optical power is: P avMAX + hcI avMAX *19 + 2.3 @ 10 *19 240 @ 10 *6 lq 1.6 @ 10
+ 20 log
Thus the optical dynamic range, DO is: DO = PavMAX - PavMIN = -4.6 -(-29.4) = 24.8dB.
In order to calculate the optical dynamic range the incident optical power must be considered.
1995 Apr 26
12
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
This represents the maximum limit attainable with the NE5210 operating at 200MHz bandwidth, with a half mark/half space digital transmission at 850nm wavelength.
APPLICATION INFORMATION
Package parasitics, particularly ground lead inductances and parasitic capacitances, can significantly degrade the frequency response. Since the NE5210 has differential outputs which can feed back signals to the input by parasitic package or board layout capacitances, both peaking and attenuating type frequency response shaping is possible. Constructing the board layout so that Ground 1 and Ground 2 have very low impedance paths has produced the best results. This was accomplished by adding a ground-plane stripe underneath the device connecting Ground 1, Pins 8-11, and Ground 2, Pins 1 and 2 on opposite ends of the SO14 package. This ground-plane stripe also provides isolation between the output return currents flowing to either VCC2 or Ground 2 and the input photodiode currents to flowing to Ground 1. Without this ground-plane stripe and with large lead inductances on the board, the part may be unstable and oscillate near 800MHz. The easiest way to realize that the part is not functioning normally is to measure the DC voltages at the outputs. If they are not close to their
+VCC 47F C1 C2 .01F
quiescent values of 3.3V (for a 5V supply), then the circuit may be oscillating. Input pin layout necessitates that the photodiode be physically very close to the input and Ground 1. Connecting Pins 3 and 5 to Ground 1 will tend to shield the input but it will also tend to increase the capacitance on the input and slightly reduce the bandwidth. As with any high-frequency device, some precautions must be observed in order to enjoy reliable performance. The first of these is the use of a well-regulated power supply. The supply must be capable of providing varying amounts of current without significantly changing the voltage level. Proper supply bypassing requires that a good quality 0.1F high-frequency capacitor be inserted between VCC1 and VCC2, preferably a chip capacitor, as close to the package pins as possible. Also, the parallel combination of 0.1F capacitors with 10F tantalum capacitors from each supply, VCC1 and VCC2, to the ground plane should provide adequate decoupling. Some applications may require an RF choke in series with the power supply line. Separate analog and digital ground leads must be maintained and printed circuit board ground plane should be employed whenever possible. Figure 4 depicts a 50Mb/s TTL fiber-optic receiver using the BPF31, 850nm LED, the NE5210 and the NE5214 post amplifier.
GND
R2 220 C9
D1 LED 1 2 100pF 3 4 R3 47k 5 6 7 8 9 LED CPKDET THRESH GNDA FLAG JAM VCCD VCCA GNDD TTLOUT IN1B IN1A 20
C7 100pF 19 C8 10 0.1F 11 12 13 14 8 9 GND GND GND GND VCC VCC NC 7 6 5 4 3 2 1
L1 10H
R1 100
C5 1.0F
C4 .01F
C3 10F
.01F
NE5210
CAZP 18 CAZN 17
C6
IIN NC GND GND
OUT1B 16
OUT GND OUT
L2 10H C11 .01F
NE5214
BPF31 OPTICAL INPUT
IN8B OUT1A IN8A RHYST
15 14 13 12
C10 10F
L3 10H
C12 10F
C13 .01F
10
RPKDET 11
R4 4k VOUT (TTL)
NOTE: The NE5210/NE5217 combination can operate at data rates in excess of 100Mb/s NRZ The capacitor C7 decreases the NE5210 bandwidth to improve overall S/N ratio in the DC-50MHz band, but does create extra high frequency noise on the NE5210 VCC pin(s).
SD00330
Figure 4. A 50Mb/s Fiber Optic Receiver
1995 Apr 26
13
Philips Semiconductors
Product specification
Transimpedance amplifier (280MHz)
NE5210
1 GND 2
14 OUT (-)
2
13 GND 2
GND 2
3
12
OUT (+) NC
INPUT
4
11
GND 1
NC 5
10
GND 1
VCC1 6 9
GND 1
ECN No.: 06027 1992 Mar 13 VCC 2 7 8 GND 1
SD00488
Figure 5. NE5210 Bonding Diagram carriers, it is impossible to guarantee 100% functionality through this Die Sales Disclaimer process. There is no post waffle pack testing performed on Due to the limitations in testing high frequency and other parameters individual die. at the die level, and the fact that die electrical characteristics may shift after packaging, die electrical parameters are not specified and Since Philips Semiconductors has no control of third party die are not guaranteed to meet electrical characteristics (including procedures in the handling or packaging of die, Philips temperature range) as noted in this data sheet which is intended Semiconductors assumes no liability for device functionality or only to specify electrical characteristics for a packaged device. performance of the die or systems on any die sales. All die are 100% functional with various parametrics tested at the wafer level, at room temperature only (25C), and are guaranteed to be 100% functional as a result of electrical testing to the point of wafer sawing only. Although the most modern processes are utilized for wafer sawing and die pick and place into waffle pack Although Philips Semiconductors typically realizes a yield of 85% after assembling die into their respective packages, with care customers should achieve a similar yield. However, for the reasons stated above, Philips Semiconductors cannot guarantee this or any other yield on any die sales.
1995 Apr 26
14

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